Thursday, December 30, 2021

An SDR For All Seasons

 

 Random Internet screen shot of Vintage Lock-In Amplifiers

I recently acquired one of the Open Source, Analog Devices Pluto Software Defined Radios (SDR) [1] to do some experiments on making a DC-1 GHz Lock-In Amplifier out of it. Since I had some questions when starting it up and I saw that the Internet had some confusion also, I thought I would add my findings and research.
 

Figure 1 – My Pluto connected to my trusty, 45-year-old HP355D, 100 dB Attenuator, to simulate the loss of a test sample in my initial Lock-In Amplifier investigations.

The Analog Devices Wiki for the Pluto [2] does a great job of getting you the right drivers, etc. They also have a demonstration program called “IIO Scope” which is a quick way to get the device running. They also have videos on the web that go through a lot of this.

I suggest that you follow the directions from the Wiki [2]. I run Windows 7 / 10 and I had no issues with any of the drivers or IIO Scope.


Frequency Extension

[Edited  9Jan22 - This section works for all Pluto's so far. If you have a Rev D PCB and want the frequency extension AND access to the other two RX/TX channels, skip ahead to the next section.]

The first issue that comes up is that Analog Devices is somewhat unclear about modifying the software configuration to extend the frequency range from 325-3800 MHz to 70-6000 MHz (Figure 2). They kind of lead you the believe that this configuration modification is only available for the very first devices that they made (Figure 2). Upon further research, it turns out that even the current Matlab Toolbox does this configuration upgrade on every device, and it does indeed work on my Rev C software, Rev D PCB (The current design as of Dec 21).

Figure 2 – The note from the Analog Devices Wiki is unclear as to the ability to do the frequency range extension configuration change for all Pluto’s – but it does indeed work for ALL Pluto's ever made (as of Dec 2021).

The exact commands used for the Frequency Extension Configuration Change are also somewhat muddled on the Wiki, but for the current Rev C / Rev D PCB Pluto the commands shown in Figure 3 are all you need. To log into the Pluto I just used the serial terminal that I use for embedded debugging as the Pluto enumerates itself as a Serial COM port also. See the Device Manager in Windows to find the COM port where the Pluto has been enumerated as. The default user name is: “root” and the password is: “analog” (without the quotes). Again this is all on the Analog Devices Wiki.
 
Figure 3 – Log in to the Pluto with your favorite terminal program and type the commands as shown above and like magic you will have an extended frequency range Pluto. (Rev C / Rev D PCB running FW Version 0.32, these commands may change in the future, this is current as of Dec 2021).

You can use IIO Scope to confirm that the device will now sample at 61.44 MHz and tune to 6 GHz as shown in figure 4.

Figure 4 – By running IIO Scope you can confirm that the configuration changes now work. The Properties panel now will let you set the bandwidth, sampling rate, and upper-frequency range to the enhanced limits as shown.

 

Possibly Useful Configuration Changes

[Edited- 9Jan22 - This section is for Rev C Pluto's with Rev D PCB's only]
The Pluto has a single RX and TX port to the outside of the box via SMA connectors, but the AD9363 transceiver chip that the Pluto is based on has two RX and two TX ports. The Rev D PCB has the extra two ports pinned out to UFL snap connectors that are populated on the board (inside the case). There is a configuration command that activates the firmware in the Pluto to be able to access these two extra ports. The command is,

fw_setenv mode 2r2t

However, there is a problem with the Firmware Version 0.32 that my Pluto arrived with. According to a post on the Analog Devices User Forums [3], there is an issue that prevents the 2r2t command from actually surviving a reboot. So it won't work.

The post describes that you should download [4] the latest firmware zip file, yes the entire zip file, place it in the root directory of the Pluto's onboard drive and then follow the rest of the procedure to upgrade the firmware as detailed here [5]. Today (January 2022) the latest version of firmware is 0.34, that is what I used.

With FW version 0.34 to get both the extended frequency range and the 2 RX and 2 TX paths enabled use these commands,

fw_setenv attr_name compatible
fw_setenv attr_val ad9361
fw_setenv compatible ad9361
fw_setenv mode 2r2t

Then reboot the device. So far this has worked for me - Now I have both extended frequency range and both RX / TX ports enabled.

If you have problems and the settings get confused, just re-update the flash to get everything to a known state and try again. If things go badly, you can always revert to the version that your Pluto shipped with.

Firmware updates always scare me, but I updated my Pluto about ten times today trying different versions out and they all went just fine. So just follow the instructions here [4] and it will all (probably) be fine.

To change back to 1r1t mode just use this command then reboot the Pluto,

 fw_setenv mode 1r1t

Note: Setting the 2r2t mode reduces the maximum sampling rate from 61.44 MSPS to 30.72 MSPS.


Useless Configuration Changes

There is a lot of 'chatter' on the internet on turning on the second ARM core in the ZYNQ FPGA that is used in the Pluto. No one pushing this change can point to any real benefit, however. 

Remember, the XC7Z010 FPGA is not a multicore Intel i7 processor, but instead has two seperate high-performance ARM cores that can indeed share peripherals, but they aren't "multicore" in the same sense as a modern Intel i7 processor that can 'spawn' threads quite easily onto other cores. Hence the standard Pluto firmware won't use the other core, just because you turn it on.

Analog Devices 'official' response is: Is it won't do any good on the standard Pluto (See below). If you are going to write custom FW for the Pluto then, yes by all means you can use the second core as you wish. All that turning on this core will do for the standard Pluto is again 'presumably' turn on the second ARM cores clocks and probably consume more power without any benefit to the standard Pluto.

 


If you read or watch any of the Analog Devices Webinars, then Robin will be a familiar name to you. Here is his take on turning on the second ARM core on the stock Pluto. Bottom Line: It won't help you at all. Source: SignalsEverywhere Youtube channel.

The Analog Devices Wiki documents all the firmware configuration options that may be set - see Reference [6] below.


RF Performance

So the next question that arises is: “What is the performance”, well I have not tested the EVM yet, but just doing a CW source to Single-frequency receiver FFT shows that the uncorrected response is perfectly fine for what this device was designed for, there is no undue roll-off in the extended frequency bands, so excellent job Analog! (See Figure 4).

 

Figure 4 – Using the freeware program SATSAGEN [7] getting a source to receiver plot is easy. This plot, made without any corrections shows a very flat source to receiver response curve over the full 70-6000 MHz frequency range is just over 2 dB peak-peak. I did use a 10 dB attenuator between the TX and RX ports on the Pluto to improve the match.

I also measured the match of the RX and TX ports on my Network Analyzer as shown in Figures 5 and 6. The match is also perfectly acceptable for what this device is intended for.
 
Figure 5 – S-Parameter Plot of the RX port match over the full frequency range. This is perfectly usable for what this device was intended for.
 
Figure 6 – S-Parameter Plot of the TX port match over the full frequency range. This is perfectly usable for what this device was intended for.


Software Compatibility

I found that the Pathosware open-source collection of tools [8] was the easiest way to get GNU Radio and the Pathos Flow programs running on a Windows PC. The only gotcha that I found was that the current release needs Python for Windows 3.9.0 installed. The current Python release is 3.10 and this did not work for me. This information is current as of December 2021 and will change in the future, so be sure to check the Pathosware release notes. I had issues getting the SDR radios to show up with every other release of GNU Radio for Windows install that I tried, but Pathos ware worked the first time out of the box. This is probably because Pathosware uses its own SoapySDR radio drivers. Pathosware also has excellent tutorials on GitHub along with instant help (I got a question answered the same day).


Programming With Other Languages

It looks like Analog Devices has a pretty robust binding package for Python, but my go-to language is C#. Analog Devices also has C# bindings on the Github page [9] and I was able to build a working example in an hour without any major issues, that’s a first, I assure you! Just start a .NET Framework project using a Winform or Console project, and add all the *.cs files from the Analog Devices Github c# bindings to your project. You don’t have to add any references to any DLL’s as Windows will find the proper DLL's at runtime.

On my first compile, I got the error as shown in Figure 6. I searched for this error and found a Forum Post at Analog Devices that told me that this was because I had my C# program in “Any CPU" mode and that the DLL that is referenced is a 64 bit DLL. So by changing the program to "X64" mode this error went away.
 

Figure 6 – If you get this error from the Analog Devices C# example program, then switch to “X64” mode instead of “Any CPU”. The reason is that the DLL being called here is a 64 Bit DLL and won’t work in 32 bit, "X86" mode.

The only other error that I found was with the initial context instantiation in the supplied Example Program. The sample IP address shown in the Example Program is not correct.

You can address a locally connected Pluto one of three ways,

1) Through the USB descriptor (USB:3.7.5) (Your exact digits will be different, here and will change every time the Pluto is plugged into a different USB Port on your PC, so beware).
2) Through the default Pluto IP Address (ip:192.168.2.1)
3) Through the string: “ip:pluto.local”


To verify these Addresses, open a Command Prompt window and type: “iio_info -s”, the results will be something like as shown in figure 7.
 

Figure 7 – The command “iio_info -s” will show you the possible descriptor strings that you can use with the C# bindings to open your Pluto. Don’t worry about the warning (see Red Arrow above), apparently this is not important, and I have even seen it in some Analog Devices Webinars and they go right past it, so it is not an issue. The important bits are the USB descriptor, shown here on my PC as: [usb:3.7.5], note however that this number will change every time you plug the Pluto into a different port of your PC, so beware. The second descriptor is the IP address shown here as “192.168.2.1”, and the third, and easiest one to use is the “ip:pluto.local”, this last descriptor never changes.

If you only have one Pluto connected to your PC then this is the way to go, just use the descriptor “ip:pluto.local” and be done with it.

If you are going to have multiple Pluto's connected then see the Analog Devices Wiki on how to change the default IP address as you will probably want to give each Pluto a unique IP address and then use that as the context descriptor.

 

Figure 8 - An excerpt from the the Analog Devices C# Example program, this context descriptor is not correct for most Pluto users, you should use instead the descriptors as shown in the list above and figure 7. The safest descriptor for most people to use is: "ip:pluto.local".


Conclusion

The Pluto uses a LINUX Industrial IO (iio) interface that also can be wrapped for windows. This is a great concept and one that works well. Overall the entire Pluto Eval Board and Software Ecosystem is very, very well thought out and well documented. Pluto works with nearly every open source SDR tool out there and is a great tool for teaching digital communications concepts. It is even is a passably good scanner receiver even without any filtering or amplification on the input.

I am running my Pluto on a very low performance Dual Core/Windows 7 computer in the lab and it works just fine, so there are not a lot of performance constraints on the host PC's horsepower either.



Extra Bonus – Filter plot

SATSAGEN has a tracking generator/receiver mode that allows a calibration by normalizing the response over a band. I dug up a 1.26 GHz Cavity notch filter that I had in the junk box and measured it with the Pluto (Figure 9). Even without optimizing anything, the dynamic range was excellent and the response showed good agreement to what my ‘real’ HP Network Analyzer showed.
 

Figure 9 – I dug up and measured with SATSAGEN an old 1.26 GHz Notch filter that I in the Lab with the Pluto. The gain response showed good agreement with what my real network analyzer measured. Thus the Pluto is a truly universal kind of RF Building Block, it even works as a Scalar Network Analyzer, and Lock-In Amplifier!

 

References:

[1] https://www.analog.com/en/design-center/evaluation-hardware-and-software/evaluation-boards-kits/adalm-pluto.html


[2] https://wiki.analog.com/university/tools/pluto

[3] User forum post on problems with 2r2t command,

https://ez.analog.com/adieducation/university-program/f/q-a/544531/pluto-rev-c-how-to-activate-the-2nd-rx-channel

[4] Pluto Firmware download location,

As per [3] download the ENTIRE Zip file, do not unzip it, just use this ENTIRE file as per the update procedure below,

https://github.com/analogdevicesinc/plutosdr-fw

[5] Pluto Firmware Update Procedure,

You will want to follow the "Mass Sorage Update" procedure listed here.

https://wiki.analog.com/university/tools/pluto/users/firmware

[6] Pluto list of firmware configurations,

See: "Updating to the AD9364" on this page,

https://wiki.analog.com/university/tools/pluto/users/customizing

See: "All Environmental Settings Table" on this page,

https://wiki.analog.com/university/tools/pluto/devs/booting


[7] SATSAGEN home page, there is no manual, but the author has how-to videos on Youtube.
http://www.albfer.com/satsagen-download-page/

[8] Pathosware homepage,

https://pothosware.com/


Pathos Project on Github,

https://github.com/pothosware/PothosCore/wiki


[9] Analog Devices Pluto C# Bindings and example project on Github

https://github.com/analogdevicesinc/libiio/tree/master/bindings/csharp

 

Post updated: 9Jan22

 

Article By: Steve Hageman www.AnalogHome.com    

We design custom: Analog, RF and Embedded systems for a wide variety of industrial and commercial clients. Please feel free to contact us if we can help on your next project. 

This Blog does not use cookies (other than the edible ones).

Sunday, November 14, 2021

The Perfect DC/DC For IoT Sensor Nodes

   


For once Battery technology and electronics are aligning pretty well for portable equipment. Sure in the last Century we could use some sort of AA Battery holder and get our supply voltage in 1.5 Volt increments, but with today's single Lithium cells operating over a 3 to 4.2 volt range and our 32 Bit Processors able to operate down to 2.5 volts, it is a far simpler task to design a small, battery-operated smart sensor node.

Sensor Operation

Sensors are naturally the heart of any remote sensing scheme. They typically run on 3 to 5 volt power supplies, and in the case of air quality sensors can draw anywhere from 30 mA to 100 mA. If your sensor node is wireless, the Radios also can draw upwards of 50 mA when transmitting.

While 100 mA at 5 Volts is a lot of power, the good news in most of these remote sensors is that they do not need to stream a lot of data. In the case of a recent Air Quality sensor project I did, it didn’t need to read faster than about 1 reading every 10 minutes, since the air quality really can’t change all that quickly. The Wireless data link and processor on time were similarly limited to just a few seconds of operation every 10 minutes, making for a very favorable power profile.

This operational profile leaves a lot of potential power optimization potential for extending battery life or reducing battery size.

Simple Sensor DC/DC

The Air Quality sensor that I was working on required two 5 Volt Sensors, an Air Particle sensor, and a CO2 sensor, the rest of the circuitry was designed to run on a continuous 2.9 Volt low power, linear regulated supply.

What is needed then to power the sensors is a 3 to 4.1 input voltage range and a 5 volt output at up to 100 mA, a further requirement of an enable control was needed to turn the sensors on only for a reading and a power good signal was required also so that the processor can judge and transmit the condition of the measuring system.

Looking around at the various manufacturer's section guides turned up the LTC1751-5, an Inductorless Switch Capacitor Voltage Doubler.

Inductorless is nice to remove the radiated EMI of a boot converters inductor from being a possible problem and possibly reduce the overall size of the DC/DC.

Figure 1 shows the basic circuit configuration of the converter. As mentioned the Power Good signal is useful for the Microprocessor to monitor if the power is good, this is an early warning that something is wrong with the sensor and this information can be sent with the wireless data back to the sensor data collection hub.

  

Figure 1 – The application circuit of the LTC1751-5 is the definition of simplicity itself. No inductors and even a Power Good output.

For instance, if the Microprocessor reads an out-of-bounds sensor value – what does that mean? But if the Microprocessor reads a out-of-bounds sensor value AND the power good signal is low, that is almost certainly a shorted sensor and the Microprocessor can not only tell the data collection node that the sensor is faulty, but the Microprocessor can also turn that sensors power off to make sure that there isn’t excessive power drain.

Application Hints

The circuit of figure 1 is nice and compact, but there are several points that are worth noting,

#1 - The output voltage ripple can be quite large – the LTC1750-5 datasheet shows 100 milli-volt peak to peak at 100 kHz with the circuit values shown. To get these values you must take care with the capacitor selection. This is one place where a ceramic capacitor's voltage coefficient can cause you trouble. Carefully look at the capacitors data sheets to make sure that your selected 10 uF output capacitor isn’t acting like a 1 uF capacitor with 5 volts of DC bias [2]. The same caution applies to C1 and C4. A couple of eye-opening articles on this subject are noted in Reference [2] below.

Capacitors of the rated value with DC Bias with low ESR are a must, and possibly secondary filtering may be required. To keep the noise levels low. See Figure 2.

#2 – The input current ripple from the LTC1751-5 running at full load is in excess of 100 mA peak to peak at 100 kHz. If operating on a small battery be sure that this amount of ripple won’t cause undue stress on the battery, especially its protection circuitry. Extra filtering may be required. A small, low ESR Super Cap may be an optimum solution to buffering the battery. Less optimum is the addition of a small LC input filter, but it can do the job very well.

#3 – Inrush, or start-up current may be a factor when running on a small battery, be sure to take this into consideration when designing the circuit. A large inrush current may cause the battery's short circuit protection to activate, especially at low temperatures.

Figure 2 – Three ceramic capacitors rated at 6.3 Volts were measured on my bench versus DC bias. The only totally safe one is the upper trace, a COG type (Blue curve above). COG types are the largest of ceramic capacitor family, but also always have a very low capacitance change with DC bias. Even X7R types can be worse than you think! Check every capacitor datasheet carefully [2].

#4 – Output noise is potentially considerable even in the suggested circuit on the datasheet, the output noise is in the 50 to 75 millivolt peak to peak range. Again you may need to increase the output capacitance and or C4 or use secondary filtering. See the datasheet for some suggestions.

#5 – Turn-on time – you have a few adjustments for turn-on time. Larger output capacitors will slow turn-on time as will adjusting the “Soft Start” capacitor (C3 in Figure 1).

#6 – Simulate or be prepared to rework: Of course, nothing beats an actual breadboard of a circuit for testing, but Linear Technology provides a “Test Jig” circuit of the LTC1751-5 for use in LTspice, it seems to simulate the noise and turn-on characteristics accurately.

#7 – If you do use an inductor-based LC filter to reduce the conducted noise, do yourself a favor and ALWAYS use a ‘Shielded’ type construction unless you want radiated noise everywhere [x].
 
Figure 3 – Linear Technology provides a LTC1751 “Test Jig” circuit for use in their LTspice circuit simulator, it is highly recommended that you simulate your circuit before committing to an actual PCB. Do be sure to accurately model the capacitor ESR - because this default Test Jig From Linear Technology uses ideal capacitor models, that's kind of overly optimistic for any power supply! Power Supply Circuits Live and Die based on the capacitors ESR!

Alternatives

You might be thinking: “Why not just use a regular Boost DC/DC Converter, it has a built-in LC filter on the input that reduces input current ripple, so it might be simpler in the long run.” See Figure 4.

 

Figure 4 – A standard Boost DC/DC converter has the power inductor on the input circuit that reduces the input ripple current, and that can simplify the overall application circuit. But it does have a potentially major fault for a battery-operated circuit, and that is it has no inherent short circuit protection. If the output is shorted for whatever reason, the controller can do nothing abort it as there is a straight-through path from the input to the output through the inductor to the boost diode to the short circuit.

And that is a valid point, but in my battery application, the short circuit protection was a requirement to keep the main application running in case of an Air Quality Sensor failure. That short circuit protection combined with the “Power Good” signal out of the LTC1751-5 gave me a level of onboard diagnostics to be able to deal with a potential sensor failure.

Bonus Information

Some manufacturers have wonderful online/interactive tools that allow you to pick any of their ceramic capacitors and get a display of all the pertinent parameters including ESR and the effect of DC Bas on the capacitance value.

One such tool is from the AVX Website [4], and picking a typical 10uF, 6.3V, X7R, 0805 capacitor produces this result,

 


Figure B1 – A typical Capacitance value drop of a 10 uF, 6.3V, X7R 0805 capacitor that took only seconds to generate from the AVX website.

 

Figure B2 – A typical Capacitance value drop of a 10 uF, 6.3V, X7R 0805 capacitor that took only seconds to generate from the Murata website.

Beware #1 – This is typical data only – i.e. Even the manufacturer didn’t go measure each, and every one of their thousands of different capacitors.

Beware #2 – Don’t think that all manufacturer's capacitors will perform the same as these will, while they might be similar, they won’t be the same and you could still expect to see variation from these capacitors from some other brand. If there isn’t an interactive tool, then you will just have to search the datasheets, or better yet, make your own measurements.


References

[1] Linear Technology LTC1751 Data Sheet

[2] Mark Fortunato, Maximum Integrated Circuits Tutorial, TU5527,
https://pdfserv.maximintegrated.com/en/an/TUT5527.pdf

Also this excellent article,
https://passive-components.eu/dc-and-ac-bias-dependence-of-mlcc-capacitors-including-temperature-dependence/

[3] Hageman, "Friends don't let friends use un-shielded inductors",

https://analoghome.blogspot.com/2017/10/friends-dont-let-friends-use-un_5.html


[4] I am sure that these links will be broken in just a few months from now, but it will be on the AVX and Murata websites somewhere, just go there and search for them....
 

AVX Online Selector Tool,

https://spicat.kyocera-avx.com/mlcc
 

Murata online tool, they call it ‘SimSurfing’
https://ds.murata.co.jp/simsurfing/mlcc.html?lcid=en-us

These sorts of tools from manufacturers are always changing - be sure to check all the manufacturers websites for the latest available information. 


Article By: Steve Hageman www.AnalogHome.com    

We design custom: Analog, RF and Embedded systems for a wide variety of industrial and commercial clients. Please feel free to contact us if we can help on your next project. 

This Blog does not use cookies (other than the edible ones). 

Monday, October 11, 2021

Preventing Design and Support 'Oopsies'

    

This is a story of how easy it is to have ‘Oopsies’ in the development of products, new, and old. It’s all related to the Data Sheet, and mostly out of your control.

Case 1:

Over a decade ago new FPGA designs came online and naturally everyone used one since they were 'big and fast'. In one FPGA project I was associated with we could not get the FPGA to ‘turn over’, this was perplexing. Everything looked wired OK, yet the FPGA would not start. Since everything looked to be done according to the data sheet, what was the issue? Everyone started to take an interest in this and another engineer downloaded the datasheet directly from the manufacturer's website ‘freshly’ as it were.

Well – ‘Big Surprise’ !!!! This new revision of the datasheet had added in the fact that the power supplies were required to be sequenced in a certain way for proper operation, where the previous datasheet said nothing of the sort! Once the Power Supplies were sequenced as per the latest datasheet – then the FPGA started right up.

Now you know why there was suddenly a rash of FPGA power supply sequencing IC’s released a decade ago!

Lesson learned: At the start, middle, and end of the current project, download ALL the critical parts datasheets and check them for changes (at least check the revision date). IC manufacturers are rushing just as fast as everyone else, and you have no idea when they will find an ‘issue’ and report it.

Aside: This is probably the reason why so many so many FPGA designers are Bald, they didn’t start out that way, it just comes from scratching their head so much.
 

Why FPGA Designers are mostly bald! 

Photo by Ketut Subiyanto from Pexels, used with permission.


Case 2:

Last year I started working on an Air-quality Data Logger. This was a battery-powered device with an LCD and SD Card for data logging. I started the project and immediately found that the I2C connection to the Air-quality sensor was not reliable at the rated 100 kHz clock speed. But seemed to work at 40 kHz and below, well to measure Air Quality doesn’t take Gigabits of bandwidth, so I started running reliability tests at 40 kHz, by sending random commands and making random measurements and found that there were no connection issues. I have used the I2C with this particular hardware setup for years with no issues at all, so I doubted that the issue was on my end. The fact that I could not reach the datasheet speeds was still disconcerting, but you know – put off till later.

As things go this project got put aside for more pressing projects, and about 6 months later I had time to start it up again. I found the same issue and I found that if the Air quality Sensor hung a power cycle was required to clear it. The operation mode was to have the air quality sensor off for 10 minutes then turn it on for one minute to make a measurement then turn it off again, so this seemed like a sure-fire way to clear any errors that may happen in real use.

Since it had been six months I had in the meantime forgotten what the sensors units of measure were, so on a summer afternoon break under a big Oak Tree, I downloaded that datasheet again on my Tablet to refresh my memory….. and – well you know what I saw! Yes, the datasheet had been updated the previous month to show that a non-standard I2C delay needed to be inserted between the writing of the command byte and the subsequent reading of the data. Normally an I2C compliant device would use “Clock Stretching” to achieve the required delay, but…

When I got back to the lab, I modified my I2C driver and – Viola! No communication issues even at 400 kHz!

Lesson learned: Don’t ignore issues like this, at the very least call the manufacturer or see if they have a user forum, or search the web for some forum entries on the part to see if anyone else has found the issue you have. If I hadn’t had the project break, I would have not found this datasheet change and would not have had the confidence to know if the driver was robust or not.

Aside 1: This is not the first time that I have had a timing issue between sending a command to some device and having some issue with how long it takes the sensor to process the command and send the result. So hopefully in the future, if I see this problem again I will hook up my logic analyzer and vary the delay between the command to return to see if the issue can be resolved as experience has shown this is the most likely issue.

Aside 2: The most common issue with I2C devices is the Stop/Start versus Repeated Start operation, while in theory, most devices like EEPROM’s work equally well with either a Stop/Start or Repeated Start sequence, many modern and complex IC’s do not. Also, many data sheets are unclear as to their particular device's exact requirements. So if you can’t get your device to respond correctly try exchanging a Stop/Start with a Repeated Start and visa versa.

  

STM32 I2C Register Configuration Bits


Case 3:

I have used this one microprocessor for several project generations, so there is little risk when updating one generations product to the next, and that is mostly limited to the changed parts of the design that happens between generations. I was wrapping up the Firmware Drivers on one of the latest projects and out of the blue, I decided that I should check the latest Errata on the part to see if anything got better. Sometimes, just sometimes, Silicon changes for the better between revisions, after all.

As I read the newest Errata, which had only been updated the month before, I was aghast to see that the required processor wait states had increased for the newest silicon revision! This doesn’t affect any parts already built, but certainly does affect any new parts that might be built, and the scary part is it was by total happenstance that I found the issue to begin with. A simple fix for this latest project, but I needed to go ‘touch’ all the previous Firmware on any shipping product to roll this fix it. That entails doing at least some targeted QA on all the projects to look for unintended side effects also. This could be a large unplanned cost.

Think about what can happen here: Someone designs a product, it works, then they move onto something else, and as long as it keeps working, they never look back, yet the part changed its characteristics in the meantime, which can now cause ‘corner case’ failures where there were none before it. The issue is: no one is in charge of ‘Maintenance’, so if someone doesn’t notice this change you can end up with a lot of iffy parts in the field that fails for baffling reasons, under certain conditions.

Lesson learned: Check the Datasheet for Errata at the beginning and end of every project to make sure that you understand the changes that have happened, don’t be lulled by ‘Familiarity Bias’. Don’t rely on the manufacturer issuing a PCN (Product Change Notice), because everyone has a different ‘definition’ of what a PCN is and this change did not elicit that kind of trigger from this manufacturer.

Aside 1: Sometimes Erratas are buried in the pages of the data sheets – most Microprocessors have 500-page plus data sheets and who can get through that? So look at the end of the datasheet where the Revision History is kept and try to discern when and where the changes have been made.

   

Microchip dsPIC Silicon Revision Eratta

Bottom Line:

A small number of manufacturers, and even some distributors allow you to sign up for ‘product updates’ – Always do this. Sadly not enough do, also sadly most manufacturers don’t use PCN’s in the spirit that they were intended: “Any change to form fit or function that might cause a customer grief”.

The only thing you can do is to keep checking the datasheets for updates, especially when actively working on a project. It is a less than ideal situation, but a very important one.



Article By: Steve Hageman / www.AnalogHome.com
We design custom: Analog, RF, and Embedded systems for a wide variety of industrial and commercial clients. Please feel free to contact us if we can help with your next project.

Note: This Blog does not use cookies (other than the edible ones).




Saturday, September 25, 2021

The Digital Filter You May Not Have Known You Were Using…

 

(Random Screen Grab From The Internet)

You use this digital filter knowingly or not, every day – it is inherently built into every Digital Volt Meter (DVM) by the fact that a common DVM is set to integrate its input signal over one line frequency cycle on every measurement. Most people know that the integration period of 1/60 Hz (or 1/50 Hz depending on where you live), or one power line cycle, allows the DVM to reject the 50 or 60 Hz noise and that noise is everywhere. You simply can’t get away from line frequency voltages and interference in the modern world. It turns out that this natural line frequency rejection is a form of a digital filter and it follows the Sine X / X, or Sinc() function as shown in Figure 1.

 


Figure 1 – Sine X / X, or Sinc() function frequency response. This response is ‘naturally’ built into every DVM by their integration over an exact number of power line cycles. This sampling then provides a null at the power line frequency, providing rejection to any power line noise. This is perhaps the most common form of a Digital Filter Around. The Blue Curve above is the Sync() filter frequency response, the Orange lines are the asymptotes that converge on the -3 dB response point.

The Sync() function is equivalent to sampling the input voltage with a rectangular window of time T.

The great advantage of the Sync() function is a Null at known intervals, and smooth transient response, this is of great advantage in a DMM. Its disadvantage is the passband is not very flat and has a long, droopy roll-off as is characteristic in these linear phase sort of filters.

As figure 1 shows the -3dB point is at 0.45 times the first null frequency. Table 1 lists some other pertinent roll-off points. The Frequency is normalized to the first null frequency.

Table 1 – A Sync() filter has various amplitude roll-off points as listed in the table above. The frequency is normalized to the first null frequency. Example: A DMM integrating for 1/60 seconds would have a -6 dB noise bandwidth at 36 Hz (60Hz*0.61 = 36Hz).

This same filter function happens if I take a 1 Mega-Sample Per Second (MSPS) ADC and store it, then average the samples over a 1/60 Hz (16.6 milli-second) period. Essentially the ‘integration time’ is 1/60 Hz and the same Sinc() filter function is again the result.

The integration time of the DVM can also be thought of as the ADC’s ‘Aperture Time’. That 1 MSPS ADC that was used in the averaging example also has an ‘Aperture Time’ that is the length of time that its internal Sample and Hold takes to switch from: Acquire to Hold modes. Aperture time, as the Analog Devices tutorial on Sample and Hold Amplifiers [1] points out,

“Perhaps the most misunderstood and misused specifications are those that include the word aperture. The most essential dynamic property of an is its ability to disconnect quickly the hold capacitor from the input buffer amplifier. The short (but non-zero) interval required for this action is called aperture time.”

Yes, misunderstood and in modern ADC’s not even commonly specified on the datasheet! The Acquisition Time is commonly specified as the time required to acquire any signal to the specified accuracy, but the Aperture Time itself is not.

Most higher speed ADC’s now have a very, very, very small Aperture Time, this can be inferred by the Analog input bandwidth which is usually much, much greater than the ADC's Nyquist frequency. For instance: A Linear Technology LTC2328-16, 16 Bit, 1 MSPS ADC has a specified analog input bandwidth of 7 MHz. The Data Converters have these large bandwidths to accommodate under-sampling and Nyquist folding of the input signals. In the LTC2328 case, this input bandwidth limitation is due to the Analog Input Circuit inside the ADC, not the Aperture Time. As an example calculation on the upper limit of Aperture Time on this ADC let's assume that the input bandwidth is set by the Aperature time and not the RC time constants in the ADC’s input circuit.

From Table 1 the -3dB point is found to be 0.45 the first null frequency, so if the -3dB point is 7 MHz, that would mean that the Aperture Time of this ADC is: 7 / 0.45 = 15 MHz or 1/15 MHz = 66 nSec. This is entirely wrong of course as the input bandwidth is not set by the Aperture Time, but by the input circuits time constants. This is however an upper limit on how big the aperture time could be to explain the input bandwidth effect, meaning: “The Aperture Time could not be longer than this and still get the rated input bandwidth of 7 MHz”

Analog Bonus Circuit

It is possible to simulate the Sync() filter with an OPAMP active filter circuit. Philbrick / Nexus research published this analog Sync() filter way back in 1968 and was named after its inventor, Henry Paynter who was a partner in the firm [2].

This circuit of figure 2 is a Third Order Liner Phase Shift ‘Paynter’ filter. Paynter filters are still widely used today in filtering biological types of signals and are even synthesized in many ‘digital’ implementations.

As per Philbrick: “...This filter is particularly useful for averaging functions of non-stationary random variables since its transient-response time is minimum for a given averaging time…” [3]

 


Figure 2 – The circuit for a Paynter filter that simulates a Sync() function response. All the values are normalized to the ‘notch frequency’.

Notch is at Wc = 1/(RC) or Fc = 1/(2*PI*R*C) Hz


 


Figure 3 – Frequency Response of the Paynter filter of figure 2 (Orange Curve) with the ideal Sync() filter response superimposed on it (Blue Curve). This filter was designed to have the 'Null' at 60 Hz.

 


Figure 4 – Linear phase filters like the Paynter have good transient response without excessive overshoot as shown from this LTSpice simulation of figure 2.

References:

[1] Analog Devices “Sample-and-Hold Amplifiers” MT-090
[2] https://en.wikipedia.org/wiki/Henry_Paynter
[3] Application Manual for Operational Amplifiers
December 1965, Philbrick / Nexus Research, Dedham, MA


Article By: Steve Hageman / www.AnalogHome.com
We design custom: Analog, RF, and Embedded systems for a wide variety of industrial and commercial clients. Please feel free to contact us if we can help with your next project.

Note: This Blog does not use cookies (other than the edible ones).

Saturday, September 19, 2020

How Common Mode Currents Are Generated in Switched Mode Power Supplies


Any Switched Mode Power Supply (SMPS), DC/DC converter, or any topology, be it Push-Pull, Single-Ended or Resonant will produce common mode currents or currents that flow between the input and output that are common to both the power rails and the ground terminals [1]. Figure 1 shows in simplified form a, typical single-ended SMPS that will be used to show how the common-mode currents arise.
 
Figure 1 –  Common-Mode currents arise because the voltage switching waveform on the transformer's primary windings causes a current to flow to the secondary side because of the transformer's inherent interwinding capacitance.

As in Figure 1, the switching action of the primary side power switch will induce a magnetic field in the transformer primary winding’s that is coupled by the secondary winding's that produces a voltage that is subsequently rectified and this is the main means of power transfer in any SMPS power supply.

However, there is also a voltage waveform that is impressed across the primary winding's and since the secondary windings are usually wound right on top of the primary [2] there will be capacitance between the input and output windings. This capacitance can be 50pF in even a small 1 Watt DC/DC converter.

When the voltage is switched on the primary there will then be a current that is induced into the secondary side via this transformer inter-winding capacitance. This current is called the “Common Mode Current” and can be measured with the circuit of Figure 2.

 
 
Figure 2 - A typical test fixture to measure common-mode currents. The current is impressed as a voltage across the 100 Ohm Resistor. The actual Current is then: Icm [Amps] = Vscope [Volts} / 100 [Ohms].

 
 

Figure 3 – Actual measured common mode current of a small 1 watt DC/DC converter of the type commonly found on cheap USB Isolators. The measurement of 18 mV p-p corresponds to 180 uA p-p.

As can be seen in Figure 3, this current is very fast and impulse like in nature and the harmonics extend well into the VHF frequency range. 18 mV peak to peak corresponds to a current of 180 uA peak to peak which is not insignificant considering that the current being common modern nature is flowing on the outsides of all the shielded cables, wires, PCB, and chassis components, etc.

The current in Figure 3 was measured from a cheap USB Isolator (Figure 5) that was meant to be inserted into a USB cable to provide galvanic isolation. And while it may do that job it will also provide a very nice VHF Impulse Generator into all of your other carefully designed circuitry.
 
Figure 5 – The Common Mode Current of Figure 3 was measured on this typical cheap USB Isolator that can be purchased for less than the price of a Large Latte. But is the cure worse than the disease?

 

Figure 6 – The common mode current now looks like what? Yes, an Impulse Noise Generator with a nice Dipole Antenna attached. All ready to put VHF noise all over your test bench.

Bottom Line:

Using an SMPS is a great way to get galvanic isolation in any circuit, but as far as Precision Analog goes it may make the “Total Noise” situation worse. As can be seen in Figure 6, adding this ‘Generator / Antenna’ into your design is probably not the desired effect that you were going to be looking for.

The only real mitigation to these Common-Mode Currents is to add inter-winding Faraday shielding inside the transformer to provide a local ground path inside the transformer for the currents or to use common-mode inductors on the input and output of the SMPS.

Perhaps a better way of providing USB isolation in instrumentation circuits is presented in the article:

https://analoghome.blogspot.com/2020/08/usb-isolation-for-instrumentation.html

References: 

[1] Over 25 years ago Jim Williams of Linear Technology did manage to find a Piezoelectric transformer that was an exciter on one end, a receiver on the other end, and a Piezoelectric bar perhaps 2 inches long in between. While this also produced common mode currents between the primary and secondary, they must have been minuscule because of the exceedingly low input to output capacitance. But this is a left-field sort of a device because of the size and price involved, which is so seldom used that it can be considered a ‘laboratory curiosity’ at best.

[2] Having any transformers primary and secondary winding's inter-wound as closely as possible reduces the Leakage Inductance between primary and secondary Leakage Inductance is where some of  the magnetic field is stored but can’t generally be used in providing useful power transfer between input and output circuits [3]. Adding Faraday Shielding between Primary and Secondary windings can reduce the common-mode currents by providing a ground leakage path, it will, however, increase the leakage inductance. Everything is a trade-off.

[3] There are always exceptions, some resonant converters are cleverly designed to utilize the leakage inductance, but these are not suited for low power designs. Again, everything is a trade-off.

 

Article By: Steve Hageman / www.AnalogHome.com

We design custom: Analog, RF, and Embedded systems for a wide variety of industrial and commercial clients. Please feel free to contact us if we can help with your next project.

Note: This Blog does not use cookies (other than the edible ones).

Wednesday, September 2, 2020

Lowering The Risk In New Designs

 

                                                           (Random Internet Screen Shot)

Gone are the days when we used to just slap some 0.1uF capacitors next to the Power Pin of an IC and be done with it. IC's are now very complex little subsystems, Take for example this cool little 18bit ADC I found. Not only does it convert at an alarmingly fast, 1 MSPS rate, but it also has a built-in amplifier/buffer with programmable voltage ranges of 2.5 to 12 Volts p-p and a low drift / trimmed reference all in one IC. The Programmable Gain Amplifier eliminates a whole lot of external signal conditioning circuitry and the 1 MSPS conversion rate makes signal processing easy because of the sheer speed of the ADC.

To do what this little 16 Pin TSSOP IC does would have taken at least 3 IC's and 1 square inch of PCB space just 10 years ago, today this is it (Figure 1).
 

Figure 1 – A complete, high-speed data acquisition system on a chip. That's it, there is nothing on the backside, and that really helps with isolation between channels. 10 years ago this would have taken at least 3 IC's and 3 times the board area. 30 Years ago and this would have been a 3x5 inch Module. The small, 5 pin IC in the upper right is a dedicated low noise regulator for the ADC's analog power requirements.

Complexity = Risk

Naturally, it has to be expected that some mistakes will be made in the design phase, either in conceptual or implementation errors. This is the fundamental trade-off in speed vs. analysis. There is a fine line that one must walk on every project.

No one wants to mess up the implementation so much so the initial breadboard does not work at all and is unfix-able. That would be a bad trade-off with too much speed. Likewise spending too much time on analysis can slow the project down and all that analysis might not uncover many conceptual errors anyway.

So what is the best way to reduce the inevitable risk of complexity? Well for starters, you can look around for other designs to leverage. To a certain degree leveraging working designs can be a great risk reducer.

On this design, however, I had no experience with the IC so what could I do? The data sheet certainly has some application information, but the real information came from the Eval Board. The good news is that with these complex IC's now the manufacturers always have to have an Eval board and these always include the schematic, parts list, and a sample layout.

With the datasheet and the Eval Board, you can at least compare what they both say. For instance, the datasheet showed the decoupling required for the internal reference section of the IC. Simple capacitors of a certain minimum size were recommended, with no mention of ESR, etc. But the Eval board told a different story. The Eval board showed slightly different values and most importantly it showed some small value resistors in series with the capacitors as if to say: "There is a minimum ESR requirement".

Assuming that the Eval board does indeed work (because the manufacturer sells it), this is good information to have. I can add the resistors to the first design and test to see if they are needed. If they aren't needed I can always replace them with zero ohm resistors going forward or remove them from the final layout. That is much easier than hacking in resistors to an already built board. What a drag that is!
 

Figure 2A – The datasheet said to bypass the Reference Pins like this.

 

Figure 2B – The Eval board says a slightly different story than the data-sheet (Figure 2A). I chose to follow the Eval board because it is known to work whereas the data-sheet is more: "In Theory". After all, it is easier to replace a few low-value resistors with jumpers than it is to add resistors to a board that does not have them.

Don't Believe Everything That You Read

The Eval board probably has a Material List and that is also a great start. Look up the parts and study their performance, but don't believe everything. In precision Analog Signal Processing you want to use COG dielectric capacitors in the signal path, this included the reference circuit(s). COG types not only have a very low-temperature drift, but they also have almost no capacitance drop with DC bias and most importantly they have no mechanical noise. This 'Mechanical Noise' is often overlooked and it has to do with the piezoresistivity [1] of the ceramic material. Many capacitor dialectics are Piezoelectric, that is if you induce stress onto the capacitor, it will produce a small voltage on its own. This can sadly often be found by simply tapping a built PCB with a pair of tweezers, or flexing the board, even slightly. As you can well imagine this undoes all the hard work of even the most carefully designed analog circuit.

For power supply rails we can use X7R types, these capacitors have higher Capacitance-Volume (CV) ratio density than COG but have no piezoresistivity effects. X7R types also have less dramatic capacitance shifts with DC bias than other types and have good overall lifetimes. Finally, the COG and X7R types are routinely rated to 125 degrees C.

While there is an alphabet soup of very high CV capacitors available many types have terrible piezoresistivity and exceeding poor capacitance drop with DC voltage, they are many times only rated to 85 Degrees C. And while they are very suitable for digital circuits, all these other ceramic capacitor types should be avoided for any and all precision analog design. Don't say I didn't warn you, be very careful of the capacitor types you choose no matter what the Eval Board uses.

Bottom Line

Don't be too cautious or you won't get anything done, also don't be in too much of a hurry or your design won't work at all. Instead, be careful and check all the sources available to you to get as much implementation information as possible to ensure that you can at least get some information out of that first PCB spin.

Bonus "Non-Analog" Idea

This getting a design second opinion also works for digital IC's especially Microprocessors. Read the datasheet but also check the manufacturer's Eval Board. The great thing about Microprocessors is there are so many eval boards and prototyping boards available. For instance, if you are looking at an ST Microelectronics STM32 Microprocessor: ST Micro usually makes several Eval board for each IC and then there is Olemex, Micro-Electronica, Digilent, Addafruit, Sparkfun and other manufacturers that make boards and provide schematics and Material Lists to compare notes with. This is an excellent way to reduce the risk when using a new microprocessor.

References

[1] https://en.wikipedia.org/wiki/Piezoresistive_effect


Article By: Steve Hageman / www.AnalogHome.com

We design custom: Analog, RF, and Embedded systems for a wide variety of industrial and commercial clients. Please feel free to contact us if we can help with your next project.

Note: This Blog does not use cookies (other than the edible ones).


Sunday, August 30, 2020

USB Isolation For Instrumentation Applications

 

USB in instrumentation applications is a 21st Century substitute for a 20th Century RS232 serial link, but with faster speeds. It still requires a conductive and grounded connection that can be several feet in length between the measuring device and the PC controller however and that is where the fun starts.

Trouble is, these communication links connect your carefully crafted measuring instrument to a PC or Laptop somewhere that may or may not have its own power supply, and then suddenly you have a massive ground loop through the AC power distribution system and all the other test instruments that may be in use.

Additionally now, nearly every test instrument includes a switching power supply that also produces ground currents due to the common-mode currents generated across their internal transformers. While these common mode currents can be carefully controlled and minimized by adding electrostatic shielding inside the transformer's windings, profit pressures have made such shielding very unlikely now.

This problem is created by common-mode currents generated by switching power supplies running down all the cables and completing the loop through the AC Mains connection. This loop is always many feet in diameter. Remember that a loop many feet in diameter has quite a large inductance and voltage across an inductor is proportional to the current rate of change,

           Vinductor = L * dI/dT

Even 60 Hz leakage current has a rate of change and will react with an AC Mains loop to produce unwanted 60 Hz voltages on your test bench. While this is not a concern and seldom noticed when probing around an embedded processor circuit. Having several switching power supplies in test instruments and the controller PC, etc. produce real havoc with low-level measurements.

Many times you can see the effects of this AC Mains ground loop by moving your hand around and touching various pieces of test equipment, the large capacitance of your body will change the path of the ground loop and you can observe the measurements changing. This is most readily apparent in real-time when looking at Oscilloscope displays when moving your hand about but it can be seen with sensitive DMM's also in that the DC values appear to change as you move your hand around.


What to do?

While running a noise-sensitive process or measurement sequence, the test instruments like a DMM, Oscilloscope, or Logic Analyzer can be disconnected from the measuring circuit, the controller PC remains, and that ground connection from the PC and it's switching power supply.

With a Laptop, the charger can sometimes be disconnected to cut that ground loop there, but this will only work for testing a few hours at best until the Laptop battery runs dead. This is not a solution for any tests, such as encountered when averaging very low noise measurements that can take 24 hours or more to run.

How about isolating the USB using one of those cheap USB isolators? There are many little isolators available for purchase online that do indeed cut the galvanic loop in the USB cable, but these common isolators also introduce noise of their own. This is because they add isolation in the USB digital side and any downstream device needs at least some trickle power to properly function. To get this trickle power these devices use – yes, you guessed it a very low cost and oftentimes low-quality DC/DC converter. These converters lack the necessary shielding on their isolation transformers and hence inject yet another source of noise inline, this time on the USB cable where you want it least.

These noise currents have to go somewhere and they will usually end up flowing back to each other through the AC Mains. That is again a huge ground loop.

 

Figure 2 – A generic USB Isolator based on the very popular Analog Devices ADuM3160 USB Bus Isolator chip [1]. This isolation works on the USB side and still contains another noise generator, namely in a very low-cost DC/DC converter (Black Block at Top of PCB) that is generating its own common-mode noise between the isolated system grounds. The Capacitance between the isolated grounds of this unit was measured at 44 pF.


Measuring the common mode current of the cheap Isolator from Figure 2 reveals that the DC/DC converter produces 18mV p-p across 100 ohms (180 uA p-p) as can be seen in Figure 3.
 

 

Figure 3A – The little DC/DC converter that is typical of these designs is not shielded and produces a lot of common-mode current. Using the test circuit shown in Figure 3B, it is measured as 18 mV p-p across 100 Ohms or 180 uA p-p. As can be seen the noise is impulse like and thus produces harmonics will into the VHF region where any wire length will act as an antenna launching the noise all over your lab.

 

Figure 3B – Typical test circuit for measuring the common mode noise of a DC/DC converter. Running from a battery source and a restive load not grounded to anything else and placed on a metal ground plane bonded to all the test instruments improves the measurement of the true common-mode noise. The voltage measured on the scope is converted to current by the formula: Icm = Vcm/100.

The other worry of these low-cost isolators is what isolation do they really have? Are they even Hipot tested? My suggestion is that if you want to use one of these that you use one that uses the full Analog Devices chipset like the Adafruit 2107 Isolator [2] because it uses a quality DC/DC converter that is Hipot tested by Analog Devices. Yes, it costs 5x more than the cheap designs, but at least you know the quality of the components.


Another Way

There is an alternative way to isolate the USB and that is to do the isolation on the Microprocessor side. This can be built into a system from the start. Since many projects use FTDI like USB to UART bridges [3] to translate the USB to a UART compatible serial signal, an optically coupled Isolator can be used between the FTDI chip UART output and the Microprocessor UART input. This has the advantage of not needing yet another DC/DC converter and that extra noise that that will produce.

A suitable Isolator is the OnSemi FOD8012A [4] as shown in figure 4. This part is rated for 15 Mbit/Sec operation which is well matched to the speed of the FTDI232RL Bridge IC. Normally I run my designs at 115,200 Baud for applications that require only low speeds to 921,600 Baud for systems that need more speed and native Windows Baud Rate compatibility to upwards of 3 M Baud for systems that can use a nonstandard Windows rate but need maximum throughput. The FOD8012A is well suited for these tasks.

 


Figure 4 – The FOD8012A is a simple to use 8 pin device. No additional DC/DC converter is required for operation since the USB side power can be supplied by the existing USB connection and the Host side is supplied by the existing Microprocessor power. Diagram provided by the courtesy of OnSemi.

 
Figure 5 – A complete circuit diagram of the "other, quieter" way to get USB isolation for a measurement product. The FOD8021A is simply added between the FTDI FT232RL USB Bridge chip and the Microprocessor Serial port lines. The FOD8021A can even do voltage translation from a 5 Volt to 3.3 Volt system.

 
Figure 6 – The actual implementation tucked into the corner of a PCB. The White FOD8012 can be seen as straddling the two ground regions. The measurement between the USB and measuring circuit ground shows 10 pF of capacitance between them which could be decreased by further by increased separation of the isolated and instrument ground regions on the PCB.

One thing to remember is that the outer metal shell of the USB connector cannot be connected to chassis ground or this will defeat the isolation. In the design idea presented here I use a 3D printed plastic bushing that I put in a larger hole than normal through the instrument's chassis that still is a tight fit around the USB connector shell. Doing this adds the mechanical support to the chassis necessary to keep the USB connector from being ripped off the PCB, while still maintaining the integrity of the ground isolation by keeping separation of the chassis ground and the USB connector shell.


Conclusion:

By isolating the USB connection after the USB/UART bridge IC another source of noise is eliminated and the overall system noise problems will be simplified. The bonus? Only one low-cost IC is required.
 

Bonus:

More information on how Common Mode Currents are generated in power supplies is available in this article,

http://analoghome.blogspot.com/2020/09/how-common-mode-currents-are-generated.html


References:
[1] Analog Devices ADuM3160 USB bus side isolator.
https://www.analog.com/en/products/adum3160.html

[2] Adafruit USB Isolator Product 2107
https://www.adafruit.com/product/2107

[3] FTDI Chip, USB to USART bridge chips.
https://www.ftdichip.com/Products/ICs/FT232R.htm

[4] OnSemi FOD8012AHigh Speed Digital Opto-Isolator
https://www.onsemi.com/products/optoelectronics/high-performance-optocouplers/low-voltage-high-performance-optocouplers/fod8012a


Article By: Steve Hageman / www.AnalogHome.com

We design custom: Analog, RF and Embedded systems for a wide variety of industrial and commercial clients. Please feel free to contact us if we can help on your next project.

Note: This Blog does not use cookies (other than the edible ones).